A New Audio Amplifier Circuit Design. The problem with class-B amplifier design is that we start with an output stage in two halves, each with a non-linear response, which we then add togethe
A New Audio Amplifier Circuit Design.
The problem with class-B amplifier design is that we start with an output stage in two halves, each with a non-linear response, which we then add together to try to give a linear response, i.e. so that a graph of output voltage vs. input voltage is a straight line.
The term "complementary symmetry" is often mentioned in connection with class-B output stages, as if there were some advantage in symmetry. Symmetrical responses of the two halves only guarantees that when added together the resulting non-linearity is also symmetrical.
To achieve a linear response what we need are two non-linear responses which add up to give a straight line. The simple solution to this problem is to start with a single half of the output stage with a conventional non-linear response, and then subtract this response from a straight line to give the response needed for the other half of the output stage.
To achieve this we can start with the well known circuit shown next:
This circuit gives an output across its load equal to the difference between its inputs. This arrangement, with the addition of a power transistor, is then used as the top half of the class-B circuit shown next:
The top half of the output stage subtracts the output of the other half, V3, from the input voltage V1, which gives a voltage V2 - V0 equal to V1 - V3 across its output resistor. What we are in effect doing here is to subtract the non-linear output, V3 of the bottom half from the undistorted input signal, V1 to obtain the required response for the top half such that when we add the outputs of the two halves through the 1ohm output resistors the sum is the required straight line response. Both halves are biased onto the linear parts of their characteristics in the quiescent state by VBIAS, and so on negative half-cycles if the bottom half provides the entire output required with no error, then the top half will not have any change in its output and will remain operating at constant current and not be cut off as in a conventional class-B circuit. On positive half-cycles the bottom half will eventually cut off, and then the top half provides the whole output current.
Some may recognise this description as being a form of 'feedforward error correction'. The top half corrects the error from the bottom half, and remains linear at all times. The bottom half is required only to provide sufficient negative output current to prevent the top half from cutting off in an attempt to correct the error. The output is actually independant of V3 because the top half inverts this signal and adds it to the original via the 1ohm resistors to give cancellation. The only non-linearity in the circuit therefore has no effect on the output across the load. In reality of course there will not be exact cancellation, but the result is far better than conventional class-B circuits.
The resistors at the input of the top op-amp are rather inconvenient if we want to use a discrete transistor output stage. Such a stage may have a relatively low, and non-linear input impedance and the voltage drop across these resistors could then cause significant errors. Fortunately there is a better arrangement which avoids this problem, shown next:
Again, for clarity, the circuit is shown with op-amps. As before the top half inverts the output of the bottom half and adds this to the original to give cancellation of the non-linearity. The operation of this circuit was confirmed by building a practical example, which was used to obtain the distortion traces on the previous page. If instead we make the top half identical to the bottom half we obtain a more or less conventional class-B circuit, and this is what was done to obtain the distortion traces for an 'unmodified' class-B circuit. The full circuit is shown later.
This diagram shows how the currents vary in the two halves of the output stage with a sine-wave input signal. The peak output current is IP and quiescent current IQ.
Next is the circuit diagram of the practical circuit. It is a low power version, with a maximum average sine wave power output a little under 20watts. For simplicity an op-amp input stage is used.
A switch is included which in one position, as shown in the diagram, gives the improved circuit, while in the other position the amplifier becomes a fairly standard class-B circuit. A direct comparison is then possible. Initial tests at 1kHz were unhelpful because there is sufficient negative feedback to reduce the distortion below the noise level of the test equipment used for any quiescent current above 10mA, so the measurements were repeated at 20kHz to give the distortion traces shown previously. An 8ohm load was used, and a 100mV 20kHz sine wave input test signal. The 20kHz sine-wave harmonic distortion is of course not going to be audible, but with a complex music signal components around 20kHz could produce intermodulation products further down the frequency range which could then be audible, so high frequency linearity is still important.
The distortion measuring equipment used a signal nulling technique in which the input signal is added to an attenuated version of the inverted output signal to give nulling of the undistorted signal, leaving the distortion alone visible. This avoids the need for a very low distortion test signal, and also avoids any phase distortion of the distortion waveform observed. The equipment used is capable of measuring distortion components over 140dB below the signal level when used with a narrow bandwidth wave analyser, and actually compares the voltage between the output terminals with that across the input terminals, to ensure any earth line distortion effects are not missed. This method requires very precise adjustment of gain and phase to give adequate nulling, but the results are worth the effort.
The distortion observed for the improved circuit was no better than the standard circuit at low quiescent current, e.g. 6mA, but as the current is increased distortion falls to a minimum at 10mA for the standard circuit, and then rises sharply again above this current. The modified circuit also falls to a low level at 10mA, but the big difference is that increasing the current further reduces the distortion further, so that it rapidly fell below the noise level, and remained low for any higher current. Precise setting of current is no longer needed. The standard circuit was difficult to adjust for a precise minimum distortion, and soon drifted away from the optimum setting as the amplifier warmed up in operation. The modified circuit actually has a poorer quiescent current stability, and if set to 80mA when cold, it drifted up to 100mA after a few minutes, even with thermal compensation included in the usual way with the BC184L transistor glued to the 2SB648A. A moderate drift upwards is no problem in this case, because the distortion remains low.
There is a simple way to avoid the need for any adjustment or compensation of quiescent current. Looking again at Fig. 4 it can be seen that the minimum current through the top half of the circuit is simply the value of the quiescent current. This minimum can be detected and used to control the biasing of the output stage. An example of this sort of circuit is shown here. (This needs more thought, in its present form something unpleasant could happen during clipping.) A diode in series with the top power transistor limits the voltage drop in the detection circuit, but at low currents the parallel resistor determines the voltage, and this is detected and compared to a reference voltage obtained from a diode chain. A medium power pnp transistor feeds a current into a large capacitor when the quiescent current falls below the required level, and the smoothed voltage across this capacitor controls a current source which determines the bias voltage of the output stage. Using this sort of circuit it should be possible to reduce the output resistors to improve maximum power output into low impedance loads.
In practice I don't believe this added complexity to be necessary. The relatively poor quiescent current stability compared to a standard output stage with the same value resistors is not really a problem because an exact value of IQ is not now needed. Standard circuits with resistors as low as 0.1 ohm are not unusual. There seems no good reason why the improved design should not have equally small values. If IQ varies by even +/- 50% this need not affect performance, but of course such a change in the standard circuit would cause high crossover distortion. For the present low power example the 1 ohm resistors are no problem unless you want to drive a 2 ohm load in which case they should be reduced.
A final note on component values. The resistors used were all 1% tolerance metal film. It may be difficult to obtain 1 ohm or 1/2 ohm resistors of 1% tolerance, and these resistors also need to be rated at 2 watts or more. Those used in the prototype were actually parallel combinations of four 3R9 0.6watt 1% metal film. The exact values of the two output resistors are not important, only their equality. The component values shown are actually not theoretically correct for accurate nulling of distortion because the lower 1 ohm resistor is in parallel with two 100 ohm resistors in series, and so more current is fed to the output via these resistors. A 200 ohm resistor connected in parallel with the upper 1 ohm resistor would correct for this, but the error is less than the tolerance of the components, so fairly unimportant. The inductor in the output circuit is to reduce the effect of capacitive loads on loop stability. I used 17 turns, 8mm diameter and 15mm length.
The Burr-Brown OPA604 was chosen for its low distortion of 0.0003% at 1kHz, low open-loop output impedance of 25R, maximum voltage rating of +/- 24V, and a 20MHz gain-bandwidth. In this circuit output current is only taken in one direction, so the class-B output stage of the op-amp is of little importance.
Only one prototype has been built, and the stabilisation circuit in Fig. 6 has not been tried at all, so I can give no assurance that even using the same type of components the same performance will be achieved. The only problem I would expect to encounter with different components is the stability margin of the feedback loop. Some experimentation with the value of the 1p8 compensation capacitor may be necessary. The power transistors used were just whatever I had available at the time I built the prototype, so they are unlikely to be the best possible choice. If other types are tried the voltage, current and power ratings do of course need to be adequate. The output 'triples' involve a feedback loop round three transistors, and this is potentially unstable. Without detailed analysis my guess is that stability is more certain if the first two transistors of the triple are fairly fast (Ft 100 to 200MHz) and the power output device relatively slow, e.g. 5MHz or less, so that this gives a dominant phase lag. The excellent results obtained with what was just a first attempt, put together in a hurry, suggests that component types are not highly critical. As always, good layout is important, avoiding large current loops, particularly in the output stage, using single point earthing, and so on.
There is one problem with this particular circuit, which is that the voltage drop across the 100 ohm feedback resistors in the output triples will become significant at high power levels, and may become sufficient to cut off the upper half during high negative output currents. This depends on transistor current gains, load impedance etc, and could be reduced by reducing the feedback resistors. In the low power version presented here the effect is not too serious, but for a high power version a different output stage becomes desirable. I have several possible designs, and may add one to this website if I ever find the time to build and test one.
If the 'less than 20W' power rating is not enough it is possible to keep the existing design by building two amplifiers and making one non-inverting so that they can be used as a bridge amplifier. A power rating well over 50 watts should be obtainable by this means. A simple switching arrangement could enable a 20 watt per channel stereo amplifier to become a mono bridge amplifier with higher power, giving the option of starting with one low power stereo amplifier and later adding another to upgrade to two higher power bridge amplifiers.
Only one prototype amplifier has been built so far, and no detailed investigation carried out to determine the effects of component variations, so it is not presented as a final design suitable for construction by the inexperienced.